Low complexity frequency domain echo canceller for DMT transceivers

ABSTRACT

A low-complexity frequency domain echo canceller for use in a full-duplex DMT transceiver. The transceiver includes a modulator that utilizes DMT modulation methods for communication over the transmission medium, a demodulator for receiving DMT signals, and a frequency domain echo canceller for removing echoes of the transmitter&#39;s signal from the receive signal. The echo canceller models the echo channel characteristic as an FIR filter that distorts the transmitted signal that is received by the receiver section of the full-duplex transceiver. The echo is removed from the received signal by determining the frequency domain characteristic of the echo channel. A local replica of the echo is generated in the frequency domain using the echo characteristic and is subtracted from the frequency domain representation of the received signal.

BACKGROUND OF THE INVENTION

A. Field of the Invention

The present invention relates to communication transceivers. Moreparticularly, the invention relates to an echo canceller designed to beoperated within a DMT full-duplex transceiver.

B. Description of the Related Art

1. Discrete Multi-Tone Modulation

Discrete Multi-Tone (DMT) uses a large number of subcarriers spacedclose together. Each subcarrier is modulated using a type of QuadratureAmplitude Modulation (QAM). Alternative types of modulation includeMultiple Phase Shift Keying (MPSK), including BPSK and QPSK, andDifferential Phase Shift Keying (DPSK). The data bits are mapped to aseries of symbols in the I-Q complex plane, and each symbol is used tomodulate the amplitude and phase of one of the multiple tones, orcarriers. The symbols are used to specify the magnitude and phase of asubcarrier, where each subcarrier frequency corresponds to the centerfrequency of the “bin” associated with a Discrete Fourier Transform(DFT). The modulated time-domain signal corresponding to all of thesubcarriers can then be generated in parallel by the use of well-knownDFT algorithm called Inverse Fast Fourier Transforms (IFFT).

The symbol period is relatively long compared to single carrier systemsbecause the bandwidth available to each carrier is restricted. However,a large number of symbols is transmitted simultaneously, one on eachsubcarrier. The number of discrete signal points that may bedistinguished on a single carrier is a function of the noise level.Thus, the signal set, or constellation, of each subcarrier is determinedbased on the noise level within the relevant subcarrier frequency band.

Because the symbol time is relatively long and is followed by a guardband, intersymbol interference is a less severe problem than with singlecarrier, high symbol rate systems. Furthermore, because each carrier hasa narrow bandwidth, the channel impulse response is relatively flatacross each subcarrier frequency band. The DMT standard for ADSL, ANSIT1.413, specifies 256 subcarriers, each with a 4 kHz bandwidth. Eachsub-carrier can be independently modulated from zero to a maximum of 15bits/sec/Hz. This allows up to 60 kbps per tone. DMT transmission allowsmodulation and coding techniques to be employed independently for eachof the sub-channels.

The sub-channels overlap spectrally, but as a consequence of theorthogonality of the transform, if the distortion in the channel is mildrelative to the bandwidth of a sub-channel, the data in each sub-channelcan be demodulated with a small amount of interference from the othersub-channels. For high-speed wide-band applications, it is common to usea cyclic-prefix at the beginning, or a periodic extension appended atthe end of each symbol to maintain orthogonality. Because of theperiodic nature of the FFT, no discontinuity in the time-domain channelis generated between the symbol and the extension. It has been shownthat if the channel impulse response is shorter than the length of theperiodic extension, sub-channel isolation is achieved.

2. Asymmetric Digital Subscriber Lines

Asymmetric Digital Subscriber Line (ADSL) is a communication system thatoperates over existing twisted-pair telephone lines between a centraloffice and a residential or business location. It is generally apoint-to-point connection between two dedicated devices, as opposed tomulti-point, where numerous devices share the same physical medium.

ADSL is asymmetric in that it supports bit transmission rates of up toapproximately 6 Mbps in the downstream direction (to a subscriber deviceat the home), but only 640 Kbps in the upstream direction (to theservice provider/central office). ADSL connections actually have threeseparate information channels: two data channels and a POTS channel. Thefirst data channel is a high-speed downstream channel used to conveyinformation to the subscriber. Its data rate is adaptable and rangesfrom 1.5 to 6.1 Mbps. The second data channel is a medium speed duplexchannel providing bi-directional communication between the subscriberand the service provider/central office. Its rate is also adaptable andthe rates range from 16 to 640 kbps. The third information channel is aPOTS (Plain Old Telephone Service) channel. The POTS channel istypically not processed directly by the ADSL modems—the POTS channeloperates in the standard POTS frequency range and is processed bystandard POTS devices after being split from the ADSL signal.

The American National Standards Institute (ANSI) Standard T1.413, thecontents of which are incorporated herein by reference, specifies anADSL standard that is widely followed in the telecommunicationsindustry. The ADSL standard specifies the use of DMT modulation.

SUMMARY OF THE INVENTION

There exists a need for a reduced complexity echo canceller that allowsthe full spectrum to be used in both directions in a symmetriccommunication system.

A low-complexity frequency domain echo canceller for use in afull-duplex DMT transceiver is provided. The transceiver includes amodulator that utilizes DMT modulation methods for communication overthe transmission medium, a demodulator for receiving DMT signals, and anecho canceller for removing echoes of the transmitter's signal from thereceive signal. The echo canceller models the echo channelcharacteristic as an FIR filter that distorts the transmitted signalthat is received by the receiver section of the full-duplex transceiver.The echo is removed from the received signal by determining thefrequency domain characteristic of the echo channel. A local replica ofthe echo is generated in the frequency domain using the echocharacteristic and is subtracted from the frequency domainrepresentation of the received signal.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other objects, features and advantages of the presentinvention will be more readily appreciated upon reference to thefollowing disclosure when considered in conjunction with theaccompanying drawings, in which:

FIG. 1 depicts a preferred embodiment of the communication system;

FIG. 2 shows a preferred embodiment of the DMT transceiver;

FIG. 3 shows a timing diagram of frames transmitted and received by theDMT transceivers; and

FIG. 4 shows a flowchart of the echo cancellation method.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The transceiver described herein is used in a communication system asshown in FIG. 1, which consists of transceiver 10, transceiver 20, andcommunication medium 30. One transceiver is located at the subscriberlocation, typically a residence or business office, and the other islocated at a central office location, or service provider location. Thecommunication medium may be a standard land-line connection over twistedpair cable or may be a wireless service between the DMT transceivers.The transceivers 10, 20 are substantially identical and are shown inFIG. 2.

With reference to FIG. 2, the transceiver of the preferred embodimentincludes a transmitter portion 200 and a receiver portion 250.Transceivers 10, 20 use wide-band multi-carrier modulation known asdiscrete multi-tone (DMT), where each channel is split into a number ofsub-channels, each with its own carrier. Preferably the number ofcarriers is two hundred fifty six, but the transceiver is easilyscalable to use additional (or fewer) carriers. The data bits are mappedto frames of complex frequency domain symbols and transformed digitallyusing a frequency-domain to time-domain transformation on each frame. Adiscrete Fourier transform (DFT) provides a computationally efficientimplementation of such a transformation.

Alternatively, one of many well-known wavelet transformations may beused to generate a modulated time-domain signal. In such a case, theinformation symbols are modulated onto a family of wavelets where eachwavelet occupies a different frequency range. Typically, each wavelet isa time-scaled version of the other wavelets in the family such that thewavelets are orthogonal. Typically, the wavelets also occupy differentbandwidths, with, e.g., the longer wavelets occupying the smallerbandwidths at the lower frequency bands, and the shorter waveletsoccupying larger bandwidths at the higher frequencies. In this sense,the wavelet transformer also results in a multi-carrier signal similarto a DMT signal, with each wavelet acting as a separate “carrier”.

The transmitter 200 of either of the transceivers 10, 20 includes a datasource 202 that provides scrambled data to modulator 204. By scramblingthe data, source 202 ensures continuous data transitions. In the MODmodulator 204, the data is mapped to signal points chosen from aconstellation of complex signal points. The IFFT transformer 206performs an inverse Fourier transform on the complex points to generatea time-domain sequence. The periodic extension is appended in theperiodic extension block 208 to the signal to allow for channel impulseresponse and to enable receiver symbol timing recovery and clocktracking. The modulator 204 or the IFFT transformer 206 also scales theamplitude of the digital signal according to the range of the D/Aconverter 210, and the data is sent through the hybrid 212 across thechannel 30 to the receiver portion 250 of the distant transceiver

The data source 202 scrambles the data to de-correlate it such that theenergy of the time-domain transmit signal is spread evenly across thespectrum. This also ensures a proper peak-to-average signal. A suitablescrambler algorithm is that used in the V.series modems, specificallyITU Recommendation V.34. The performance and complexity of thisalgorithm are well known, and code exists for its implementation oncommon DSP platforms. Alternatively, a block based scrambler using alookup table may be used.

The MOD modulator block 204 maps input data to complex points in asignal constellation for each sub-channel. One of a number of modulationtechniques may be used. Quadrature Amplitude Modulation (QAM), MultiplePhase Shift Keying (MPSK) (including QPSK), Differential Phase ShiftKeying (DPSK) (including DQPSK) and the like, are all possiblemodulation schemes. DQPSK is presently preferred. The data is mapped asa phase transition. At the receiver the phase of each carrier iscompared to its previous phase from symbol to symbol. This has theadvantage of resolving phase ambiguities between the transmitter andreceiver.

In the presently preferred embodiment, modulator 406 is a 256 tone DMTmodulator, based on a 512 point IFFT. The analog bandwidth of the signalwill depend upon the D/A conversion rate used in the particularimplementation. The sample rate may be, e.g., 16 MHz, and in accordancewith the well-known Nyquist sampling theorem, this implies that thetotal possible usable bandwidth is the range from 0-8 MHz.Alternatively, a subset of the 256 carriers across the 8 MHz bandwidthmay be used by specifying a magnitude of zero for any carriers not used.

To generate a real-valued time-domain signal using an inverse DFT, a 512point Inverse Fast Fourier Transform (IFFT) is performed, where the last256 points are reverse-ordered complex conjugates of the first 256points. It is a well-known property of discrete Fourier transforms thatreal-valued time domain signals have conjugate-symmetric Fouriertransforms.

Voice-band frequency content may be eliminated in the modulated signalby setting the frequency bins corresponding to the voice-band to zero.The modulated DMT signal would therefore not interfere with POTS devicesoperating over the same channel. Similarly, the frequency binscorresponding to any other data services present on the LAN medium maybe set to zero.

The formula for the IFFT inverse transform is:${d_{j} = {\frac{1}{\sqrt{n}}{\sum\limits_{k = 1}^{n - 1}{w_{k}^{{- 2}\pi \quad {{(\frac{j}{n})}}k}}}}},$

for 0≦j<512, where the d_(j) are the time domain data points, n is thelength of the IFFT, w_(k) are the complex-valued symbols, and i={squareroot over (−1)}. The w_(k) are set to a zero value for binscorresponding to frequencies that are not used. The above summationbegins at k=1 because w₀ is preferably zero.

The periodic extension or prefix is appended in block 208. The periodicextension, or cyclic extension as it is often referred to, is arepetition of the beginning samples of the time-domain signal generatedby the DFT and is appended to the time-domain signal. One of ordinaryskill will recognize that a periodic or cyclic prefix is an equivalentto the periodic extension. The length of the periodic extension ispreferably at least as long as the impulse response of the channelbetween the transmitter of transceiver 10 and receiver of transceiver20. The model of the channel impulse response includes echoes fromunterminated wiring segments that may be present within the medium 30(e.g., in a business or residential environment having numerous wiringruns). The length of the periodic extension is computed based on theworst-case channel impulse response time, the worst case expectedreflected echo tails, and the expected symbol (frame) timing errorencountered at the receiver. The periodic extension must also be longenough to accommodate echo cancellation. The transmit signal fromtransceiver 10 is received by its own receiver, together with manyechoes of the transmit signal, which appear as overlapping, scaled, andtime-shifted versions of the transmit signal.

Once the time-domain transmit signal is generated for the current frameof data, the D/A converter 210 transmits it to hybrid 212 fortransmission over the full-duplex communication link. The receiver 250of FIG. 2 receives signals from the distant end as well as echoes of anysignals transmitted by its own transmitter 200.

In normal operation the receiver must first obtain framesynchronization. Receive frame synchronizer 254 performs this functionby the use of a correlator to detect the repetitive samples of theperiodic extension. Alternatively, a frame synchronization method usingpilot tones may be used. In such a scheme the phases of two adjacentpilot tones transmitted in the first frame are compared to determine theframe index. Because a sampling offset results in a progressive phaseoffset from bin to bin of a DFT, an examination of the extent of thephase offset between two known symbols will yield the sampling offset,and thus the frame index. The cyclic prefix is removed (or in the casewhere an extension is used, the beginning of the frame is replaced withthe extension), and the data samples representing the data circularlyconvolved with the channel are sent to transformer 256.

Transformer 256 then performs a transform of the real valued time domainsignal and generates a complex frequency domain signal. The first framecontains known data and is used to determine the equalizer coefficientsin the equalizer 260. Equalizer 260 processes subsequent blocks ofreceived data using these coefficients and updates the coefficientsbased on an error signal generated by error vector calculator 266.

Transformer 256 operates on the synchronized time-domain data togenerate the frequency domain spectrum. The Fourier transform usedwithin block 256 takes the real valued time-domain receive samples thathave been properly framed and produces an output consisting of complexvalues containing real and imaginary components. The function used isequivalent to:${w_{j} = {\frac{1}{\sqrt{n}}{\sum\limits_{k = 1}^{n - 1}{d_{k}^{{- 2}\pi \quad {{(\frac{k}{n})}}j}}}}},$

for 0≦j<n, where the d_(k) are the time domain data points, n is thelength of the FFT, w_(j) are the complex-valued symbols, and i={squareroot over (−1)}.

The transformed frequency domain data represents the magnitude and phaseof the carriers. The FFT points are commonly referred to as “frequencybins.” The length of the output will contain half as many points as thereal valued time-domain receive signal because only the first half ofthe points are calculated. As stated previously, the other half of thefrequency domain points are merely complex conjugates of the desiredpoints, and are therefore not needed.

The data is then equalized in block 260 and demodulated in block 262.The equalizer 260 is a frequency domain complex equalizer thatsimultaneously solves the problems of symbol timing error, clock errorand drift, channel phase and attenuation distortion, and removes anynumber of echoes caused by reflections of unterminated wiring segments.This is accomplished in one mathematical step of low complexity.

The transmitted data may be arranged in packets, with each packetconsisting of a limited number of concatenated frames transmitted inserial fashion, or may be arranged as a continuous stream of frames. Ineither case, the initial frame is an equalization frame of known symbolsthat is used to provide a coarse estimate of the channel. The receiver'sequalizer 260 is trained to the channel using this frame by forming theratio of the expected symbol to the received symbol for each frequencybin within the frame. The ratios for each tap of equalizer 260 areformed using the complex-valued frequency domain values of the receivedsignal passed through the channel. Those values are readily availablefrom the FFT block 256. The result is a sequence of points (e.g., avector) where each point corresponds to a frequency bin, and each valueis an estimate of the inverse of the channel response at that frequency.

Multiplication of the output of FFT module 256 by the equalizer tapsresults in a circular de-convolution of the channel impulse response.The circular de-convolution is made possible by the periodic extension,which makes the receive data to appear as if it had been circularlyconvolved by the channel impulse response. Thus the single step ofmultiplying the transformed data frames by the equalizer coefficientsprior to demodulation corrects for channel impulse response distortion,sampling offset, clock/timing error, etc.

A decision feedback loop is used after the demodulator 262 to generatean error vector in block 266 that is used to update the equalizer tapsafter each frame is processed. Block 266 allows the equalizer to trackslow changes in the channel and to track clock error between thetransmitter and receiver.

The demodulator block 262 takes the complex frequency domain points foreach bin after equalization, then demodulates those points back to realdata. Demodulator 262 includes data slicers to determine the nearestconstellation point to the received (and equalized) point. Thedemodulator may include a trellis decoder and other forward errorcorrecting decoders. Data module 264 reverses the scrambling performedby the transmitter section based on the V.34 scrambler, or a block-basedlookup table.

In the full-duplex transceiver, the transmit echo is removed in a mannersimilar to the way channel distortion is removed from the receivedsignal. The echo is modeled as the transmit signal passing through anFIR filter that is then received by the receiver. The echo canceler 270removes the echoes by determining the frequency response of the FIRfilter. This is possible because the cyclic extension is as long as theecho impulse response, and is therefore long enough to allow the echocharacteristic to be determined.

Upon transmission of a single frame containing training symbolsT(w_(j)), and in the absence of a signal from the distant endtransmitter, the echo signal is received and processed by A/D converter252, frame synchronizer 254 and time-domain to frequency-domaintransformer 256. The resulting frequency domain points U(w_(j)) arecompared to the original frequency domain points T(w_(j)) provided bymodulator 204. The ratio of T(w_(j))/U(w_(j)) is the echo characteristicEC(w_(j)), and comprises a single coefficient for each frequency bin, j.The echo characteristic may be better determined by transmittingnumerous training frames and averaging the ratios.

Once training is complete, the local echo signal of any transmittedframe may be replicated by passing the transmit points T(w_(j)) throughthe echo canceler. The replica is then subtracted from the receivedsignal at the output of transformer 256 via summer 258. The method isparticularly low in complexity because both receive and transmit dataare already available in the frequency domain. In addition, thesubtraction is also done in the frequency domain. None of thecomputational expenses typical of time-domain methods such asinterpolation, phase adjustments, etc., are necessary.

Transmit/receive frame synchronizer 268 ensures that transmit andreceive FFT frames are aligned. The receiver will synchronize the FFTframes to the signal received from the distant end transmitter. For echocancellation to be effective, the local echo signal must also fallwithin the frame of the received signal. If a transceiver's transmissionis perfectly aligned with the signal frame received from the distantend, then the distant end transceiver will not have aligned transmit andreceive frames due to propagation delay in both directions. Thisscenario is depicted in FIG. 3. Waveform (a) represents two frames ofdata transmitted at time T₀ from transmitter 1. Waveform (b) representsthe same two frames received at receiver 2 at time T₁. Waveform (c)represents two frames of data transmitted from transmitter 2 aftertransmitter 2 detects the signal energy from transmitter 1. Waveform (d)represents the two frames received at receiver 1. Waveforms (e) and (f)represent the composite waveforms at receiver 2 and receiver 1,respectively. The misalignment of transmit and receive frames atreceiver 1 by two-way propagation delay (and a detection delay) makeecho cancellation impossible unless a much longer cyclic prefix is usedto accommodate the misalignment.

Preferably the transceivers are frame synchronized such that bothtransmitters transmit at the same time. This imposes a less burdensomecriterion for the length of the cyclic prefix because it results in amisalignment corresponding to only one propagation time delay. This isaccomplished by a negotiation between the transceivers to measure theround-trip propagation delay and to establish coarse frame timing. Theecho characteristic EC(w_(j)) must be updated once transmit/receiveframe synchronization is accomplished by synchronizers 254, 268.Specifically, the window of samples selected for transformation bytransformer 256 is determined by receive frame synchronizer 254. Thetiming of the transmit frame is in turn determined by transmit/receiveframe synchronizer 268 partially in response to the frame synchronizer254. Any offset of the transmit echo signal within the window relativeto the window used during echo canceller training must be accounted for.Because a time offset corresponds to a progressive linear phase rotationin the bins of the transform, the echo characteristic EC(w_(j)) isupdated by performing such a rotation to each bin in response to thetime offset imposed by the synchronizers 254, 268.

The echo canceling method is depicted in the flowchart of FIG. 4. Theecho canceling method is particularly useful in a full-duplex DMTtransceiver. At step 410 the transmit echo frequency characteristic iscalculated. This is performed in the absence of a signal from a distantend transmitter, and involves transmitting a training signal andreceiving a corresponding echo signal. The echo signal is then convertedto the frequency domain and compared to the transmitted signal. Thetransmit signal is already available in the frequency domain, so a ratioof the converted echo signal to the frequency domain representation ofthe transmitted training signal is formed.

During normal operation data signals are transmitted as in step 420. Atstep 430 the composite signal consisting of a data signal from a distantend transmitter and a transmit echo signal is received for processing.The received composite signal is transformed to a received compositefrequency signal in step 440. A replica of the transmit echo signal isgenerated using the transmit echo frequency characteristic in step 450,which is then subtracted from the received composite frequency signal instep 460.

The transmit data signal in step 420 is generated by specifying themagnitude and phase of a plurality of DMT carriers and transforming itto a corresponding time-domain signal. In addition, the transmission ofstep 420 is performed in timed relation to the received data signal fromthe distant end transmitter, as determined by the synchronizers 254,268.

The replica of the transmit echo signal is formed by multiplying thefrequency domain representation of the transmit signal available fromthe transmitter 200 by the transmit echo frequency characteristic. Theecho characteristic is also updated in response to the framesychronization to account for any phase rotations imposed by the timeshifting of the echo samples.

A preferred embodiment of the present invention has been describedherein. It is to be understood, of course, that changes andmodifications may be made in the embodiment without departing from thetrue scope of the present invention, as defined by the appended claims.

We claim:
 1. An echo canceling method for use in a full-duplex DMTtransceiver comprising the steps: calculating a transmit echo frequencycharacteristic; transmitting a data signal; receiving a composite signalconsisting of a data signal from a distant end transmitter and atransmit echo signal; transforming the received composite signal to areceived composite frequency signal; generating a replica of thetransmit echo signal using the transmit echo frequency characteristic;and subtracting the replica from the received composite frequencysignal.
 2. The echo canceling method of claim 1, wherein the step ofcalculating a transmit echo frequency characteristic further comprisesthe steps of: transmitting a training signal; receiving a correspondingecho signal; converting the received echo signal to the frequencydomain; and forming a ratio of the converted echo signal to thefrequency domain representation of the transmitted training signal. 3.The echo canceling method of claim 1, wherein the step of calculating atransmit echo characteristic is performed in the absence of a signalfrom a distant end transmitter.
 4. The echo canceling method of claim 1,wherein the step of transmitting a data signal includes the steps of:specifying the magnitude and phase of a plurality of carriers; andgenerating a corresponding time-domain signal.
 5. The echo cancelingmethod of claim 1, wherein the step of transmitting a data signal isperformed in timed relation to the received data signal from the distantend transmitter.
 6. The echo canceling method of claim 1, wherein thestep of generating a replica of the transmit echo signal comprises thestep of multiplying a frequency domain representation of the transmitsignal by the transmit echo frequency characteristic.
 7. The echocanceling method of claim 1, wherein the step of transforming thereceived composite signal to a received composite frequency signalincludes performing a Fourier transform on the received compositesignal.
 8. The echo canceling method of claim 7, wherein a cyclic prefixis removed prior to performing the Fourier transform.
 9. The echocanceling method of claim 1, further comprising the step of updating thetransmit echo characteristic in response to the frame sychronization.10. An echo canceling method for use in a full-duplex DMT transceivercomprising: calculating a transmit echo frequency characteristics;transmitting a data signal; receiving a composite signal comprising adata signal from a distant end transmitter and a transmit echo signal;synchronizing the received composite signal on a receive framesynchronizer; selecting a window of samples associated with the receivedcomposite signal on the receive frame synchronizer; determining a timeoffset associated with the selected window of samples; updating the echocharacteristic based on the time offset associated with the selectedwindow of samples; generating a replica of the transmit echo signalusing the updated echo characteristic; and subtracting the replica fromthe received composite frequency signal.
 11. The method of claim 10,wherein the step of calculating a transmit echo frequency characteristiccomprises: transmitting a training signal; receiving a correspondingecho signal; converting the received echo signal to the frequencydomain; and forming a ratio of the converted echo signal to thefrequency domain representation of the transmitted training signal. 12.The method of claim 10, wherein the step of generating a replica of thetransmit echo signal comprises multiplying a frequency domainrepresentation of the transmit signal by the updated echocharacteristic.
 13. The method of claim 10, wherein the step oftransforming the received composite signal to a received compositefrequency signal includes performing a Fourier transform on the receivedcomposite frequency.
 14. The method of claim 10, wherein the step ofsynchronizing the received composite signal comprises synchronizing thereceived composite signal using a correlator.